Electronic linear scale using a self-contained, low-power inductive position transducer

ABSTRACT

An electronic linear scale, which uses an induced current position transducer, includes a slider assembly mounted adjacent to an elongated beam. The slider assembly is mounted on one mounting member of a device such as a machine tool. The elongated beam is mounted on a second mounting member of the device. The first and second mounting members move relative to each other. The position of the slider assembly on the beam is indicative to of the position between the first and second mounting members. Relative movement is determined by an inductive read head mounted on the slider assembly that couples to flux modulators on the beam. The read head includes a transmitter winding and a pair of receiver windings, preferably fabricated in a planar form, carried by a common printed circuit board. The flux modulators, which can include flux disrupters and/or flux enhancers, are carried by the beam and modulate the magnetic fields produced by the transmitter. Thus, the receiver windings produce output voltages corresponding to the overlap between the modulators and the receiver windings. Signal processing electronics are connected to the read head and determine the relative positions of the first and second mounting members by monitoring the output of the receiver windings. A digital display displays the determined position. The relatively insensitive nature of the induced current position transducer to contaminants allows the linear scale to function with improved reliability when operating in industrial environments. Low power operation of the linear scale is possible.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to an electronic linear scale, and moreparticularly, to electronic linear scales employing inductively coupledtransducer elements.

2. Description of Related Art

Electronic linear scales are common in the manufacturing industry andwherever high precision measurements are required. Linear scales areoften affixed to equipment such as drill presses, lathes, saws,numerically controlled and automated machinery. The linear scale allowsthese machines to produce goods with a high level of precision.

These scales have used transducers based on optical systems, magneticscales, inductive transformation, and capacitive devices.

Conventional linear scales have used optical technology with incrementaland absolute measurement systems. Problems with this system include arelatively high power requirement, high manufacturing cost due to theprecise mechanical tolerances required, and sensitivity to contaminants.To make a measurement, the optical system used a self-contained lampdirected at an optical scale and transmitted through (or reflected) toan optical receiver. The lamp involved in this system usually consumesmore energy than is practicably provided by conventional batteries. Thisreduces the portability and ease of installation of the linear scale.Because the lamp and the optical receiver traverse the scale during ameasurement, this linear scale requires a precise guiding track to guideand support the optical components. This track must to constructed withclose tolerances and the optical receiver must be precisely aligned, sothat the image reflected off the optical scale is focused onto theoptical receiver. Due to the close assembly tolerances required, thebearing system used in the device, and the cost of the optical-gradecomponents, the optical systems have high manufacturing costs.Additionally, the effects of wear or poor alignment significantlydegrade the performance.

Capacitive transducers draw very little current. Therefore, when used inbattery-powered measurement devices, they provide portability and easeof installation. Capacitive transducers operate under a parallel platecapacitor model. Within the capacitive transducer, a transmitterelectrode and a receiver electrode are mounted on or in a slide. Thetransmitter electrode is connected to appropriate signal generatingcircuitry. The receiver electrode is connected to appropriate readcircuitry.

The slide moves along a scale. The scale includes a plurality ofspaced-apart signal electrodes, which extend along the length of thescale. As the slide moves relative to the scale, the transmitter andreceiver electrodes on the slide capacitively couple to the signalelectrodes on the scale. The read circuitry determines the movement orposition of the slide relative to the scale. The read circuitry comparesthe phase of at least one signal coupled to a receiver electrode withthe phase of at least one signal coupled to a transmitter electrode.

The capacitive position transducer may be an incremental type transduceror may be an absolute position type transducer. In the incremental typecapacitive position transducer, the read circuitry provides only anindication of relative movement from a known point. In the absoluteposition type capacitive position transducer, the read circuitryprovides an indication of the absolute position between the slide andscale regardless of any knowledge of the initial position. Incrementaland absolute position type position transducers are disclosed in U.S.Pat. Nos. 4,420,754 and 4,879,508.

These capacitive position transducers are sometimes used in dry,relatively clean environments, such as in inspection rooms orengineering offices. However, these capacitive position transducers aredesirably usable in electronic linear scales to measure dimensions inmachine shops and other relatively contamination-filled environments. Inthese environments, capacitive linear scales can become contaminated byparticulate matter and fluids, such as metal particles, grinding dust,and cooling or cutting fluids. The liquid or particulate contaminantsfind their way between the signal electrodes on the scale and thetransmitter and/or receiver electrodes on the slide. The contaminantsalter the capacitance between the signal electrodes and the transmitterand/or receiver electrodes in a manner unrelated to the position of theslide relative to the scale. In general, contaminants between the signalelectrodes and the transmitter electrodes and/or the receiver electrodesof a capacitive position transducer cause measurement errors throughthree different mechanisms. Primarily, the particulate or liquid mayhave a dielectric constant different from the dielectric constant ofair. In this case, the capacitance between the signal electrodes and thetransmitter/receiver electrodes sandwiching the contaminant will begreater than the capacitance between other ones of the signal andtransmitter/receiver electrodes having the same relative geometry whichdo not have contaminants between them. As a result, the capacitivelinear scale will not provide an accurate indication of the position ofthe slide relative to the scale.

Secondarily, the contaminants may have a relatively high conductivity.Normally, the signal and transmitter/receiver electrodes form an opencircuit, such that no current flows between them. A conductivecontaminant between the signal and transmitter or receiver electrodescloses this circuit. In particular, the contaminant forms the resistiveelement of an RC circuit. The time constant of the RC circuit is afunction of both the conductivity of the contaminant and the capacitancebetween the signal electrode and the transmitter and/or receiverelectrodes. When the time constant is relatively short, the amplitude ofthe signal may decay so rapidly that the conventional circuitry employedin capacitive position transducers cannot properly sense the signal.

Thirdly, electrically conductive particles between the signal electrodeand the transmitter and/or receiver electrodes may alter the fieldextending between the signal electrode and the transmitter and/orreceiver electrodes. This changes the capacitance between the signalelectrode and the transmitter and/or receiver electrodes. Distortions inthe electric field may also distort the signals between the signalelectrode and the transmitter and/or receiver electrodes. As a result,the capacitive linear scale circuitry does not provide an accurateindication of the position of the slide relative to the scale.

U.S. Pat. No. 5,172,485 to Gerhard et al. describes one approach tominimizing the adverse effects of contaminants in capacitive positiontransducers. This approach comprises coating the electrodes with a thinlayer of dielectric material. The slide is then mounted on the scale sothat the dielectric coating on the slide (transmitter and receiver)electrodes is positioned adjacent to the dielectric coating on the scale(signal) electrodes. That is, placing the dielectric coatings betweenthe signal electrodes and the transmitter and receiver electrodesminimizes these adverse effects. In addition, the dielectric coating onthe slide slidingly contacts the dielectric coating on the scale. Thesliding contact between the dielectric coatings reduces the gap betweenthe slide and the scale into which the contaminants intrude.

The sliding contact approach requires that the electrodes be resilientlybiased toward each other. The resilient bias accommodates deviationsfrom exact surface flatness and alignment by permitting the electrodesto move apart from each other. This allows the dielectric layers to beforced apart from each other. Thus, when using such a capacitiveposition transducer in a highly contaminated environment, contaminantscan force the slide away from the scale, collecting between the slideand scale. Thus, this approach is inadequate under some circumstances.

However, using thick dielectric coats, rather than the thin coats taughtby Gerhard et al., has reduced, to some extent, the negative effects dueto contaminants collecting between the slide and scale. The thickdielectric coats create a pair of capacitors connected in series withthe capacitance created by the contaminants. The capacitance created bythe dielectric coats does not vary as the slide moves along the scale.Thus, the change in capacitance between the signal electrodes and thetransmitter and/or receiver electrodes resulting from changes in thethickness or the composition of the contaminants is dominated by thefixed capacitances created by the thick dielectric coats. Although usingthick dielectric coats can reduce the problem caused by dielectriccontaminants, this approach cannot completely eliminate the problem.

Another approach isolates the electrodes from the liquid and particulatecontaminants. For example, the capacitive position transducer linearscale may be sealed. However, sealing the linear scale increases thefabrication and assembly costs and is often unreliable. Also, such sealsare difficult to practically apply to all sizes and applications ofelectronic linear scales.

Magnetic transducers are alternative types of position measuringtransducers. Magnetic transducers are relatively insensitive tocontamination caused by oil, water and other fluids. Magnetictransducers, such as the Sony Magnescale encoders, employ a read headthat detects magnetic fields and a ferromagnetic scale selectivelymagnetized with one or more periodic magnetic patterns. The read headsenses changes in the magnetic field as the read head moves relative tomagnetic scale patterns on the scale. However, magnetic transducersthemselves are affected by small particles, particularly ferromagneticparticles attracted to the magnetized scale. Consequently, magnetictransducers must also be sealed, encapsulated or otherwise protected tokeep contaminants from affecting their accuracy. Magnetic transducersalso do not offer the very low power consumption desired for someapplications of electronic linear scales.

Inductive transducers, in contrast to both capacitive and magnetictransducers, are highly insensitive to cutting oil, water or otherfluids as well as to dust, ferromagnetic particles, and othercontaminants. Inductive transducers, such as the INDUCTOSYN® typetransducers, employ multiple windings on one member to transmit avarying magnetic field received by similar windings on another member.The multiple windings can be a series of parallel hairpin turns repeatedon a printed circuit board. An alternating current flowing in thewindings of the first member generates the varying magnetic field. Thesignal received by the second member varies periodically based on therelative position between the two members. A position determiningcircuit connected to the varying signal from the second member candetermine the relative position between the first and second members.However, both members are active. Therefore, each member must beelectrically coupled to the appropriate driving circuitry. This couplingis typically achieved by wired connections. When a moving wireconnection must be attached and routed for free movement, thereliability of the unit is generally lower while the maintenance costand production cost of the unit are higher. Sometimes an electricalcoupling approach based on transformers is used in place of wireconnections to the scale elements. This approach may reduce signalquality and accuracy, and may require additional size and cost toaccommodate the placement of coupling elements on the scale and readhead. Moreover, such a system would usually consume more energy than canbe practically provided with conventional batteries, reducing theportability and the ease of installation.

Other motion or position transducers that are insensitive tocontaminants are described in U.S. Pat. No. 4,697,144 to Howbrook, U.S.Pat. No. 5,233,294 to Dreoni, and 4,743,786 to Ichikawa et al., andBritish Patent Application 2,064,125 to Thatcher. These referencesdisclose position detection devices that sense position between anenergized member and an inactive or unenergized member. The transducingsystems described in these references eliminate electrical intercouplingbetween the two moving members, a drawback of inductive transducers.However, these systems generally fail to provide the high accuracy ofinductive or capacitive transducers.

Additionally, in some of these transducing systems, the inactive memberis preferably ferromagnetic, to produce a sufficiently strong magneticfield. Ferromagnetic members offer limited fabrication options, and maybecome magnetized in the presence of magnetic chucks, thereby attractingmagnetic particle. Alternatively, the inactive member is moved within amagnetic field defined and concentrated by a complex structure formed inor on the active member. The transducing systems disclosed in thesereferences also produce output signals that are discontinuous or are nota simply prescribed function of position. Such signals contribute toinaccurately determined relative positions. Generally, all of thesesystems fail to provide the combination of sufficient accuracy andextensive measuring range that is commercially demanded for linearscales.

SUMMARY OF THE INVENTION

This invention thus provides an electronic linear scale usable in harshindustrial and field environments. The electronic linear scale of thisinvention includes transducer elements that are substantiallyinsensitive to particulate and fluid contaminants. The electronic linearscale of this invention retains the conventional form and operation ofconventional electronic scales while featuring low power consumption.

The electronic linear scale of this invention is readily andinexpensively manufactured by using conventional fabrication techniques,such as printed circuit board technology. Furthermore, the transducer ofthis invention is insensitive to contamination by particles, includingferromagnetic particles, or oil, water, or other fluids. As a result,the transducer avoids using expensive environmental seals whileremaining usable in most shop or field environments. A pulse-drivencircuit of the electronic linear scale of this invention allows theinductive transducer to consume little power. Accordingly, the linearscale can derive a long operating life from a small battery or solarcell, and features increased portability and ease of installation. Inaddition, where the linear scale is powered primarily by a conventional,separate power source such as an electrical wall socket, the linearscale featuring a back-up power supply, such as a small battery or solarcell, does not require recalibration following failure of theconventional power source.

The linear scale includes a slider mounted adjacent to an elongated mainbeam in a conventional configuration. The main beam is mechanicallycoupled to an elongated scale that includes a set of magnetic field orflux modulators extending in a pattern along the length of the scale.The slider assembly includes a pickoff assembly. The read head ismounted to the pickoff assembly. The linear scale electronic systememploys electrical connections only to the read head, not to themodulators. Relative movement between the main beam and the slidercorresponds to the relative movement between the set of modulators andthe read head, respectively.

Signal processing electronics are connected to the read head. The signalprocessing electronics indicate the relative position between the readhead and the set of magnetic field or flux modulators as a function ofthe disruptive effect of the modulators on the signals produced andreceived by the read head. The electronic linear scale configuration ofthis invention preferably uses a low power inductive transducer with aread head that moves relative to the set of magnetic field or fluxmodulators. However, the electronic linear scale may also use othertypes of inductive transducers that are sufficiently accurate, usesufficiently low power, and are similarly insensitive to contaminants.

The inductive transducer of this invention includes a magnetic fieldsource preferably having a first path of conductive material. Themagnetic field source is able to produce a changing magnetic field orflux. At least one of the set of magnetic field or flux modulators ispositioned within the changing magnetic field or flux. The modulatorsspatially vary the magnetic field or flux proximate to the set ofmagnetic field or flux modulators. A sensing conductor, formed within athin zone, forms a periodic pattern of flux-receiving areas. Theperiodic pattern extends along a measuring axis and is positioned withinthe changing magnetic field or flux. The changing magnetic field or fluxthus passively generates an electromotive force (EMF) across at leastone output of the sensing conductor in response to the changing magneticfield or flux.

The at least one magnetic field or flux modulator and the periodicpattern of the sensing conductor move relative to each other from afirst position to a second position. In the first position, a firstportion of the periodic pattern overlaps the at least one magnetic fieldor flux modulator. In the second position, a second portion of theperiodic pattern overlaps the at least one magnetic field fluxmodulator. That is, the at least one magnetic field or flux modulatorvaries the EMF from the first position to the second position.

The at least one magnetic field or flux modulator cooperates with theperiodic pattern of the sensing conductor to produce an EMF amplitudeacross the output of the sensing conductor. This amplitude changes as acontinuously varying periodic function of the relative position of theat least one magnetic field or flux modulator and the periodic patternof the sensing conductor. In one embodiment, each one of the set ofmagnetic field or flux modulators is a flux disrupter formed by anelectrically conductive plate. In another embodiment, each one of theset of magnetic field or flux modulators is a flux enhancer formed froma material having a high magnetic permeability. In another embodiment,the set of magnetic field or flux modulators includes at least one fluxdisrupter and at least one flux enhancer.

The sensing conductor is preferably formed by a plurality of first loopsalternating with a plurality of second loops. The loops are formed of aconductive material. The first and second loops are positioned withinthe changing magnetic field. Each of the first loops produces a changingfirst signal component in response to the changing magnetic field.Likewise, each of the second loops produces a changing second signalcomponent in response to the changing magnetic field.

The plurality of first and second loops and the set of magnetic field orflux modulators are movable relative to each other. In a first position,one or more of the first loops may be proximate to corresponding ones ofthe set of magnetic field or flux modulators, thus altering the firstsignal components generated by those first loops. In a second position,one or more of the second loops may be proximate to corresponding onesof the set of magnetic field or flux modulators, thus altering thesecond signal components produced by those second loops. The first andsecond signal components indicate the position of each of the first andsecond loops relative to the set of magnetic field or flux modulators.

Thus, this invention incorporates an inductive sensor with advantageousphysical and electrical characteristics, accuracy, range, and powerconsumption into a practical linear scale. The linear scale isinsensitive to both particle and liquid contaminants, and is thereforesuitable for a wide variety of applications. These applications includeportable and low power applications. In addition, the linear scale isaccurate and relatively inexpensive to manufacture, assemble, andinstall, compared to conventional linear scales. These and otherfeatures and advantages of this invention are described in or areapparent from the following detailed description of the preferredembodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

The preferred embodiments of the invention will be described in detailwith reference to the following figures, wherein:

FIG. 1 is an isometric view of an electronic linear scale which containsa low power inductive position transducer;

FIG. 2 is an exploded view of the slider of the linear scale and anassociated digital display unit;

FIG. 3 is a cross-sectional view of the slider and the beam of thelinear scale taken along line a--a of FIG. 1;

FIG. 4 is a plan view showing a layout of transmitter and receiverwindings of a read head for an inductive position transducer, andcorresponding disrupter scale elements;

FIG. 5 is a plan view showing the alternating loops of one of thereceiver windings of FIG. 4;

FIG. 6 is an exploded isometric view of a second preferred embodiment ofthe invention having receiver and transmitter windings on the carrierbeam;

FIG. 7A is a plan view showing a receiver winding overlaying the scale,with the scale coupling to a first portion of the receiver winding;

FIG. 7B is a plan view showing the receiver winding overlaying thescale, with the scale coupling to a second portion of the receiverwinding;

FIG. 7C is a waveform diagram showing the output signal amplitude andpolarity, from the receiver winding as it moves versus position of thescale;

FIG. 8 is a block diagram of the encoder electronics used in the firstand second embodiments of the electronic linear scale;

FIG. 9 shows a plot of voltage versus time for a resonant signal outputfrom the signal generator;

FIG. 10A is a plot of voltage versus time for a signal output from thereceiver winding;

FIG. 10B is a plot of voltage versus time when the relative positionbetween the scale and the main beam has been moved one-quarterwavelength;

FIG. 10C is a plot of voltage versus time when the relative positionbetween the flux modulator and the receiver winding has been movedone-half wavelength;

FIGS. 11A-G are signal timing diagrams showing voltages at selectedlocations in the encoder electronics of FIG. 8 during sampling;

FIGS. 12A-G are signal timing diagrams of voltages at selected locationsin the encoder electronics of FIG. 8 where the control signal istruncated to reduce energy loss;

FIG. 13 is a circuit diagram of a second preferred embodiment of thesignal generator;

FIG. 14 is a plot of voltage versus time for a resulting signal acrossthe capacitor in the signal generator of FIG. 13;

FIG. 15 is a block diagram of the encoder electronics of the electroniclinear scale, operating in reverse compared to the encoder electronicsof FIG. 8;

FIG. 16 is a signal timing diagram showing voltages at selectedlocations in the encoder electronics of FIG. 15 during sampling;

FIG. 17 is a waveform diagram showing a plot of the voltage amplitudeout of the receiver winding versus the position of the scale;

FIG. 18 is a block diagram of encoder electronics of the electroniclinear scale, incorporating a wavelength tracker;

FIG. 19 is a signal timing diagram showing voltages at selectedlocations in the encoder electronics of FIG. 18 during sampling;

FIG. 20 is a signal diagram showing timing of signals at selectedlocations in the encoder electronics of FIG. 18;

FIG. 21 is an exploded isometric view of a third preferred embodiment ofthe electronic linear scale of this invention;

FIG. 22 is a side cross-sectional view along a line perpendicular to theline a--a of FIG. 1, of a portion of an alternative embodiment of thelinear scale showing topographically formed disrupters;

FIG. 23 is an isometric view of the scale of a fourth preferredembodiment of the linear scale of this invention, using enhancer-typemodulators;

FIG. 24 is a side cross-sectional view of the scale of FIG. 23 takenalong the line 24--24 of FIG. 23 showing the enhancers carried by thebase;

FIG. 25 is a cross-sectional view of the scale of a fifth preferredembodiment of the linear scale of this invention;

FIG. 26 is a plan view of the scale of a sixth preferred embodiment ofthe linear scale of this invention;

FIG. 27 is a side cross-sectional view of the scale of a seventhpreferred embodiment of the linear scale of this invention;

FIG. 28 is a side cross-sectional view of the scale of an eighthpreferred embodiment of the linear scale of this invention;

FIG. 29 is a side cross-sectional view of a first variation of the scaleof the eighth preferred embodiment of the linear scale of thisinvention; and

FIG. 30 is a side cross-sectional view of a second variation of thescale of the eighth preferred embodiment of the linear scale of thisinvention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

As shown in FIG. 1, an inductive linear scale 100 includes an elongatedbeam 102 and a slider assembly 120. The beam 102 is a rigid orsemi-rigid bar having a generally rectangular cross section.

During operation, the slider assembly 120 is installed in closeproximity to scale 104 without contacting any components on the beam 102or the beam 102 itself. Both the beam 102 and the slider assembly 120may be affixed, in a variety of configurations, to other objects. Inthis manner, the linear scale 100 measures the position of the sliderassembly 120 relative to the beam 102, thereby also measuring thedistance between any mechanical fixtures attached to these twocomponents.

Alternately, the scale 104 carrying modulators 170 may be fabricated ofa flexible or rigid dimensionally stable material. The scale 104 can beapplied directly to a mechanical fixture with clamps or adhesive, thuseliminating the beam 102. The scale 104 and the beam 102 are preferably,but not necessarily, non-conductive. The slider assembly 120 inductivelymonitors its own position relative to the scale 104.

The scale 104 is an elongated printed circuit board 168. As shown inFIG. 1, a set of disrupters 170 are spaced apart along the printedcircuit board 168 in a periodic pattern. The disrupters 170 arepreferably formed of copper. The disrupters 170 are preferably formedaccording to conventional printed circuit board manufacturingtechniques, although many other methods of fabrication may be used. Adisplay cable 116 is also shown in FIG. 1.

As shown in FIG. 2, the measurement generated by the linear scale isdisplayed on a conventional digital display 138 mounted on the digitaldisplay unit 139. A pair of push button switches 134 and 136 are alsomounted on the digital display unit. The switch 134 turns on and off thesignal processing electronics 166 and the digital display unit 139. Theswitch 136 resets the display 138 to zero. Alternately, the positioninformation from the slider assembly may be routed to other types ofelectronic control systems or displays. The slider assembly 120 includesa slider housing 140, which holds the read head 164 to place the readhead windings 178, 179 and 180 in close proximity to the modulators 170on the beam 102 without making contact.

The slider assembly 120 also includes a substrate 162, such as aconventional printed circuit board. The signal processing electronics166 are preferably mounted on an upper surface of the substrate 162,although a portion of the signal processing electronics may be placed onthe interior side of read head 164, as an alternative (not shown). Thecontents of the slider housing are protected by a cover 114. Aribbon-like read head connector 150 electrically connects the read head164 to the signal processing electronics 166. Display cable 116 isattached to signal processing electronics 166 inside the housing, byconventional means.

As shown in FIG. 3, the read head 164 windings are covered by a thindurable insulative coating 167. The insulative coating 167 is preferablyapproximately 50 μm thick. Alternatively, the insulative coating may bereplaced by a membrane sealing the entire scale-facing surface of theslider housing 140. All external joints of the slider assembly 120 aresealed with adhesives (not shown) or other conventional means to preventcontaminants from contacting the electronic circuits.

The slider assembly 120 carries the read head 164 so that it is slightlyseparated from the beam 102 by an air gap 174 formed between theinsulative coatings 167 and 172. The air gap 174 is preferably on theorder of 0.4 mm. Together, the read head 164 and the disrupters 170 forman inductive transducer. The inductive transducer is preferably of anysuitable type disclosed in U.S. patent application Ser. No. 08/441,769,which is incorporated herein by reference, and more particularly of thetypes described herein. However, the linear scale 100 can also use othertypes of inductive transducers that offer the necessary mechanicalpackaging attributes, are sufficiently accurate, are low power, and aresimilarly insensitive to contaminants. Power may be supplied to eitherthe slider assembly 120, or the digital display unit 138, byconventional means (not shown) and routed therebetween through thedisplay cable 116. Either battery power, power from solar cells, orconventional external power may be used. If the linear scale is batteryor solar powered, the power conservation methods described in U.S.Provisional Patent Application 60/015,707, filed Apr. 17, 1996, can beused.

FIG. 4 shows a section of the read head 164 in greater detail. The readhead 164 preferably includes five substantially coplanar conductors180-184. Two of the conductors 181 and 182 form a first receiver winding178. Another two of the conductors 183 and 184 form a second receiverwinding 179. The first and second receiver windings 178 and 179 arecentrally located on and extend along the substrate of the read head 164in an overlapped manner.

The first and second receiver windings 178 and 179 are each arranged ina sinusoidal pattern and have the same wavelength. The conductor 181extends from a terminal 185 to an interconnection terminal 189a, whereit connects to the conductor 182. The conductor 182 then extends back toa terminal 187. The conductors 181 and 182 forming the first receiverwinding 178 define a plurality of sinusoidally shaped loops 191.

Similarly, the conductor 183 extends from a terminal 188 to aninterconnection terminal 189b, where it connects to the conductor 184.The conductor 184 then extends back to a terminal 186. The conductors183 and 184 forming the second receiver winding 179 also define aplurality of sinusoidally shaped loops 192. The loops 192 are offset byone-quarter wavelength or one-half of a loop from the loops 191 formedby the first receiver winding 178.

In FIG. 4, the conductors 181-184 are shown on the same surface of thesubstrate of the read head 164. However, alternate half wavelengthsections of each of the conductors 181-184 are actually on separatelayers of the substrate. Thus, the windings 178 and 179 do notphysically contact each other. Similarly, each of the windings 178 and179 do not physically contact themselves at the "crossover" points inthe middle of the pattern. The half wavelength sections of each of theconductors 181-184 are then joined to other half wavelength sections ofthe same conductor by feedthroughs 190 extending through the substrate.While the conductors 181-184 are not on the same surface of thesubstrate, the conductors 181-184 lie within a thin zone. That is, thedistance between the topmost layer of the windings 178 and 179 on thesubstrate and the bottommost layer of the substrate is minimal.Therefore, the conductors 181-184 are approximately coplanar.

The second receiver winding 179 is substantially identical to the firstreceiver winding 178, except for the spatial phase offset. Accordingly,the following discussion will focus primarily on the first receiverwinding 178. It should be appreciated that the following discussionapplies equally to the second receiver winding 179.

The fifth winding 180 is a transmitter winding that also lies within thethin zone and substantially encircles the first and second receiverwindings 178 and 179. The transmitter winding 180 is also formed by aconductor on a layer or surface of the substrate of the read head 164.The transmitter winding 180 is also formed according to conventionalprinted circuit board manufacturing techniques. The transmitter winding180 has a length 194 and a width 195 sufficient to encircle the firstand second receiver windings 178 and 179. FIG. 4 also shows terminals197 and 198 of the transmitter winding 180.

The inductive transducer formed by the read head 164 and the disrupters170 operates, in the first preferred embodiment shown in FIGS. 1-4, bygenerating changing magnetic fields. The changing magnetic fields inducecirculating currents, known as eddy currents, in the disrupters 170placed within the changing magnetic field.

For example, one of the disrupters 170 is placed between the pole facesof an electromagnet. The magnetic field between the pole faces changeswith time, such as when the electromagnet is driven by an alternatingcurrent. Then, the flux through any closed loop in the disrupter 170will change. As a result, an electromotive force ("EMF") is inducedaround the closed loop. Since the disrupter 170 is a conductor, an eddycurrent is generated whose value equals the EMF divided by theresistance along the loop of the material from which the disrupter 170is formed.

Such eddy currents are often produced in the magnetic cores oftransformers. In transformers, such eddy currents are unwanted becausethey result in power loss and create heat that must be dissipated. Inthis invention, however, the existence of eddy currents has been appliedto provide a beneficial result.

Except as otherwise specified, measurements in FIGS. 4 and 5 are definedrelative to a measuring axis 300. "Length" generally refers todimensions extending parallel to the measuring axis 300 and "width"generally refers to dimensions extending perpendicular to the measuringaxis 300 in the plane of the substrate of the read head 164. Thedistance spanned by two adjacent loops 191 formed by the first receiverwinding 178 or two adjacent loops 192 formed by the second receiverwinding 179 is defined as the pitch or wavelength 193 of the read head164. The distance spanned by a single loop 191 or 192 is equal toone-half of the wavelength 193. The distance 302 spanned by eachdisrupter 170 is preferrably also equal to one-half of the wavelength193. The one-quarter wavelength offset between the first receiverwinding 178 and the second receiver winding 179 produces signals inquadrature. Thus, the direction of motion of the read head 164 relativeto the scale 104 is observable. Furthermore, the distance 304 spannedfrom one edge of a disrupter 170 to the corresponding edge of anadjacent disrupter 170 is preferably equal to the wavelength 193. Itshould be appreciated that if the disruptors all are substantiallyidentical, the edge-to-edge distance 304 can be any integer multiple "K"of the wavelength 193. In the later case, it is preferable that eachreceiver winding have a length of "N*K" times wavelength 193, where N isalso an integer. As shown in FIG. 5, the first receiver winding 178 hasa sinusoidal pattern of loops 191. The first receiver winding 178 isformed by the conductors 181 and 182 laid out in one direction in asinusoidal or zigzag pattern, and then in a reverse direction. Thus, theconductors 181 and 182 physically (but not electrically) cross over eachother to form the loops 191. Alternately, the loops 191 can be createdby twisting a loop of insulated wire clockwise or counterclockwise 180degrees at regular increments along the loop. The construction of thesecond receiver winding 179 is identical to the first receiver winding178.

As a result of the cross-over structure of the loops 191, adjacent onesof the loops 191 have different effective winding directions. Analternating current flowing through the transmitter winding 180 producesa uniform, time-varying magnetic field extending through the firstreceiver windings 178. The time-varying magnetic field generates an EMFin, or time-varying current through, the first receiver winding 178.Thus, the receiver winding 178 functions as a specialized magnetic fluxsensor. Since adjacent ones of the loops 191 are wound in alternatingdirections, the EMF and current generated in adjacent loops 191 havealternating polarities, as indicated by the "+" and "-" symbols in FIG.5.

Each of the loops 191 encloses substantially the same area. Therefore,if the number of "+" loops 191a equals the number of "-" loops 191b andthe loops 191 receive a uniform magnetic flux, the magnetic fieldinduces a net zero EMF across the terminals 185 and 187 of the firstreceiver winding 178. This is also true for the second receiver winding179.

If a disrupter 170 on the scale 104, or any other conductive object, ismoved close to the read head 164, the magnetic field generated by thetransmitter winding 180 will induce eddy currents in the disrupter 170or the other conductive object. Consequently, a magnetic field iscreated in the vicinity of the disrupter counteracting the magneticfield generated by the transmitter winding 180. The eddy currents thusgenerate reverse magnetic fields that attenuate the transmitter magneticfield proximate to the disrupter 170.

As a result, the first receiver winding 178 receives a spatially alteredor disrupted magnetic flux. So long as the "+" loops 191a and the "-"loops 191b are not equally disrupted , the receiver winding 178 outputsa non-zero EMF signal. Consequently, the EMF between the outputterminals 185 and 187 will change polarity as the conductive disrupter170 moves from adjacent to a "+" loop 191a to adjacent to a "-" loop191b.

The size of the disrupter 170 preferably does not equal the wavelength193. For example, assume the length 302 of the disrupter 170 is equal tothe wavelength 193 and the width of the disrupter 170 is equal to thewidth 195. Then, regardless of where the disrupter 170 is positionedalong the measuring axis 300 relative to the loops 191, it will disruptthe transmitter magnetic field over equal areas of adjacent "+" loops191a and "-" loops 191b. As a result, the amplitude of the EMF signaloutput from the receiver winding 178 will be nominally zero.

Furthermore, the output from the receiver winding 178 will beinsensitive to position of the object relative to the loops 191. Thatis, the output will be zero regardless of the disrupter's 170 positionalong the measuring axis. Since no useful signal results from thisgeometry, the size of the disrupter 170 preferably does not equal thewavelength 193. The length 302 of the disrupter 170 could be greaterthan one wavelength 193. However, because the portion of the disrupterequal to a full wavelength 193 will not contribute to the useful signalstrength, the length 302 of the disrupter 170 is preferably less thanone wavelength 193.

If the length 302 of the disrupter 170 is not equal to one wavelength193 (or integer multiples of the wavelength 193), then in mostpositions, unequal "+" and "-" areas of the loops 191 will be disrupted.The signal output will thus be sensitive to the position of thedisrupter 170 relative to the loops 191. The signal output will have alarge amplitude change as a function of position when the length 302 ofthe disrupter 170 is equal to one-half of the wavelength 193. When thelength of the disrupter 170 is one-half of a wavelength 193, thedisrupter 170 will periodically cover either an entire "+" loop 191a oran entire "-" loop 191b, but will not cover any portion of an adjacent"-" loop 191b or "+" loop 191a. Thus, a one-half wavelength-longdisrupter 170 will produce the strongest possible signal.

As shown in FIG. 4, the disrupters 170 are arranged on the scale 104with a pitch (a distance of one edge to the adjacent corresponding edge)of one wavelength 193. Thus, successive disrupters are separated byone-half of a wavelength 193. The disrupters 170 are preferably highlyelectrically conductive, but not ferromagnetic. Thus, the disrupters 170do not become magnetized and attract ferromagnetic particles. As shownin FIG. 1, in the first preferred embodiment, the length of the scale104 exceeds the length of the read head 164. Thus, the length of thescale 104 establishes the measurement range for the linear scale 100.

FIG. 6 shows a second preferred embodiment where the positions of theread head 164 and modulators 170 are reversed. The receiver windings 178and 179 (represented as cross-hatch on the read head 164) and thetransmitter winding 180 (not shown) are carried by the carrier beam 102.The disrupters 170 are part of a simplified slider assembly 121. Notethat the slider assembly 121 is rotated from its normal perspective forillustration purposes so that the disrupters 170 can be seen. In thissecond embodiment, the read head 164 and the windings extendsubstantially the length of the carrier beam 102. In addition, a signalprocessing unit 169 is electronically connected to the read head 164 onthe carrier beam 102 and a remote digital display 138. The signalprocessing unit 169 includes the signal processing electronics 166mounted on the substrate 162. In this second preferred embodiment, theread head windings, not the modulators, effectively provide the longrange position reference. In some applications, this reversedconfiguration may provide more convenient or compact installations.However, in most cases, the extended active windings should be avoideddue to the negative effects associated with unshielded externalelectrical and magnetic fields.

In either the first or second preferred embodiments of the linear scale100, the loops 191 of the first receiver winding 178 are preferablyplaced within a prescribed region in the interior of the transmitterwinding 180. The inventors have experimentally determined that thetransmitter winding 180 produces a magnetic field having an intensitythat rapidly diminishes as a function of the distance from the conductorof the transmitter winding 180. However, the inventors have alsoexperimentally determined that in the interior region of transmitterwinding 180, the magnetic field tends to approach a uniform value beyonda certain distance from the conductor of the transmitter winding 180.

The certain distance thus defines the perimeter of a region ofrelatively uniform magnetic field. The distance at which the magneticfield becomes uniform is a function of the geometry of the winding.Consequently, to improve the accuracy of the inductive transducer ofthis invention, the loops 191 and 192 are preferably spaced the certaindistance away from the transmitter winding 180. The loops 191 and 192 ofthe first and second receiver windings 178 and 179 are more preferablylocated entirely within the region of relatively uniform magnetic field.

In one exemplary embodiment, the disrupters 170, the receiver windings178 and 179, and the transmitter winding 180 are dimensioned as follows:

Receiver winding wavelength=0.200 inch;

Disrupter length=0.100 inch;

Disrupter width=0.490 inch;

Transmitter winding width=0.400 inch;

Receiver winding width=0.340 inch;

One-quarter receiver wavelength=0.050 inch; and

Transmitter winding length=1.950 inches.

By accurately balancing and alternately interleaving the "+" loops 191aand the "-" loops 191b, the first receiver winding 178 has a nominallyzero output in the absence of the disrupters 170. At the same time,locating the alternating "+" loops 191a and "-" loops 191b immediatelyadjacent to each other provides for a continuous signal at each receiverwinding output as the disrupter 170 is moved along the measuring axis300. These design factors provide a high signal-to-noise ratio in thelinear scale 100. Thus, these features enable high accuracy measurement.

The above-outlined geometry of the first preferred embodiment of theread head 164 and scale 104 ensure that the linear scale 100 is highlyaccurate. Additionally, the above-outlined geometry of the firstpreferred embodiment of the linear scale 100 largely eliminates effectsfrom non-uniform transmitter fields along the width of the read head 164perpendicular to the measuring axis 300. The above-outlined geometryalso rejects externally applied magnetic fields as "common-mode error"due to the balanced "differential detection" of the inductive transducerof this invention. The degree of accuracy in the first preferredembodiment of the inductive transducer linear scale 100 depends largelyon the care in the design and construction of the read head 164 and thescale 104.

FIGS. 7A-7C show an example of the operation of the inductive linearscale 100. As the scale 104 and its disrupters 170 (shown in dashedlines) move relative to the transmitter 180 and the first receiverwinding 178, the disrupters 170 cover either all of the "+" loops 191aand none of the "-" loops 191b, varying proportions of the "+" loops191a and the "-" loops 191b, or all of the loops 191b and none of the"+" loops 191a.

FIG. 7A shows the disrupters 170 covering all of the "-" loops 191b andnone of the "+" loops 191a of the first receiver winding 178. Thetransmitter winding 180 inductively couples to and induces eddy currentsin the disrupters 170. As a result, the disrupters 170 produce magneticfields that counteract the transmitter magnetic field passing throughthe "-" loops 191b. Thus, the net magnetic flux passing through the "-"loops 191b is less than the net magnetic flux passing through the "+"loops 191a. The "-" loops 191b therefore generate less induced EMF thanthe "+" loops 191a. Consequently, the first receiver winding 178produces a net "positive" polarity current and voltage across its outputterminals 185 and 187.

The output signal varies with time because the transmitter winding 180generates a time-varying magnetic field. The amplitude and polarity ofthe time varying output signal relative to the input signal indicatesthe relative position between the read head 164 and the scale 104. FIG.7C illustrates how the output signal amplitude and polarity vary as theposition of the scale 104 varies relative to the read head 164.

The initial peak in the waveform shown in FIG. 7C is an example of apositive polarity amplitude output across the terminals 185 and 187 ofthe first receiver winding 178. Polarity indicates the time phase of thetime-varying output signal relative to the input signal. The polarity ofthe output signal will be either in phase or inverted (180° out ofphase) relative to the input signal.

FIG. 7B shows the scale 104 moved so that the disrupters 170 overlap allof the "+" loops 191a, but none of the "-" loops 191b. In this relativeposition, the induced current generated in the disrupters 170counteracts the flux of the transmitter magnetic field passing throughthe "+" loops 191a. The "-" loops 191b thus generate more induced EMFthan the "+" loops 191a. Consequently, the first receiver winding 178generates a net negative polarity current and voltage at its outputterminals 185 and 187. The initial valley in the waveform shown in FIG.7C is an example of a negative polarity amplitude output across theterminals 185 and 187 of the first receiver winding 178.

When the disrupters 170 completely overlap the "-" loops 191b, as shownin FIG. 7A, the resulting output signal has a maximum positiveamplitude, as shown in the peaks in the waveform of FIG. 7C. Conversely,when the disrupters completely overlap the "+" loops 191a, as shown inFIG. 7B, the resulting output signal has a maximum negative amplitude,as shown in the valleys in the waveform of FIG. 7C.

As the disrupters 170 move along the measuring axis 300 between theposition shown in FIG. 7A and the position shown in FIG. 7B, theamplitude of the waveform of FIG. 7C varies continuously. In particular,the amplitude of the waveform of FIG. 7C is zero when the disrupters 170overlap exactly one-half of each of the "+" loops 191a and the "-" loops191b. From this position, as the disrupters 170 move more closely to theposition shown in FIGS. 7A or 7B, the amplitude of the receiver outputsignal is increasingly positive or negative, respectively.

The first preferred embodiment of the read head 164, as shown in FIG. 4,has two receiver windings 178 and 179 spaced one-quarter of the scalewavelength 193 apart from each other. That is, the second receiverwinding 179 overlaps the first receiver winding 178 and is offset byone-quarter of the scale wavelength 193. Thus, each "+" loop 192a of thesecond receiving winding 179 overlaps a portion of a "+" loop 191a and aportion of a "-" loop 191b of the first receiver winding 178. Similarly,each "-" loop 192b of the second receiver winding 179 overlaps a portionof a "+" loop 191a and a portion of a "-" loop 191b of the firstreceiver winding 178.

Insulation or crossover vias are suitably placed on or in the substrate164, respectively, to electrically isolate the first receiver winding178 from the second receiver winding 179. By spacing the first andsecond receiver windings 178 and 179 one-quarter of a scale wavelength193 apart, the signals from the first and second receiver windings 178and 179 are spatially in quadrature. That is, the signal amplitudesoutput from the receiver windings 178 and 179 define sinusoidal patternsas functions of position. In particular, the sinusoidal pattern of thesecond receiver winding 179 is spatially shifted 90° with respect to thesinusoidal pattern of the first receiver winding 178. As a result, thesignal processing and display electronic circuit 166 detects thetransitions between the signals from each of the receiver windings 178and 179. By analyzing this relationship, the signal processing anddisplay electronic circuit 166 determines the direction the read head164 is moving relative to the scale 104. As outlined above, theamplitudes of the signals output by the windings 178 and 179 varysinusoidally based on the position of the read head 164 relative to thescale 104. Thus, the signal processing and display electronic circuit166 determines the position of the read head 164 with respect to thescale 104 by the following equation: ##EQU1## where: p is the position;

λ is the scale wavelength 193;

n is an integer indicating the number of full wavelengths 193 traveled;

S1 and S2 indicate the amplitudes and signs of the output signalsreceived from the receiver windings 178 and 179, respectively; and

"tan⁻¹ " is the inverse tangent function defining an angle between zeroand 2π as a function of the ratio between S1 and S2. The signs of S1 andS2 define which quadrant the angle lies in according to Table 1.

                  TABLE 1                                                         ______________________________________                                        S1       S2               tan.sup.-1 (S1/S2)                                  ______________________________________                                        +        +                0 to /2                                             +        -                /2 to                                               -        -                 to 3/2                                             -        +                3/2 to 2                                            ______________________________________                                    

To improve the accuracy of the linear scale 100, and/or to reduce thedemands on the analog signal processing circuitry for the receiveroutput signal, the read head 164 can include three or more overlappingreceiver windings. While a read head 164 having three or moreoverlapping receiver windings is more difficult to manufacture, itprovides, in combination with certain signal processing techniques, moreaccurate position readings than a read head 164 having only twooverlapping receiver windings. Such multiple winding read heads arepreferably equally phase-shifted. For example, for a number of windingsm, the phase shift will be 180°/m.

FIG. 8 shows the signal processing and display electronic circuit 166 ingreater detail. The signal processing and display electronic circuit 166solves Eq. 1 and controls the electronic operation of the linear scale100. As shown in FIGS. 1 and 2, the signal processing and displayelectronic circuit 166 is mounted on the substrate 162 as a part of theslider assembly 120. Also shown in FIG. 2 is the digital display unit139, which includes the switches 134 and 136 and a conventional digitaldisplay 138. To initialize a position measurement, the signal processingelectronics 166 supply an electrical excitation signal to thetransmitter winding 180 of the read head 164 (see FIG. 8).

As shown in FIG. 8, the signal processing and display electronic circuit166 uses a programmed microprocessor or microcontroller and peripheralintegrated circuit elements. However, the signal processing and displayelectronic circuit 166 can also be implemented on an ASIC or otherintegrated circuit, a hardwired electronic or logic circuit such as adiscrete element circuit, a programmable logic device such as a PLD, PLAor PAL, or the like. In general, any device that supports a finite statemachine capable of implementing the signal processing and displayfunctions described herein can be used to implement the signalprocessing and display electronic circuit 166.

The signal processing and display circuit 166 preferably comprises amicroprocessor 226. The microprocessor 226 inputs a signal from an A/Dconverter 224. The microprocessor 226 generates and outputs controlsignals to the display 138, the A/D converter 224, a switch 225, asignal generator 200, and a delay circuit 219. The output of the delaycircuit 219 is input to the control inputs of a first sample and holdcircuit 217 and a second sample and hold circuit 218.

Each of the outputs of first and second sample and hold circuits 217 and218 are connected to one of the input terminals of the switch 225. Theoutput of the switch 225 is connected to the input of the A/D converter224. The inputs to the first and second sample and hold circuits 217 and218 are connected, respectively, to the output terminals 185 and 188 ofthe first and second receiver windings 178 and 179.

The other output terminals 187 and 186, respectively, of the first andsecond receiver windings 178 and 179 are connected to ground. The outputof the signal generator 200 is connected to the terminal 197 of thetransmitter winding 180. The other terminal 198 of the transmitterwinding 180 is also connected to ground.

To perform a position measurement, the signal processing and displayelectronic circuit 166 supplies an electrical excitation signal to thetransmitter winding 180 of the read head 164.

As shown in FIG. 8, the first sample and hold circuit 217 has a bufferamplifier 216 connected to one of the terminals of the switch 225. Theoutput terminal 185 of the first receiver winding 178 is connectedthrough a switch 221 to the input of the buffer amplifier 216. Thecontrol terminal of the switch 221 is connected to the delay circuit 219and inputs the sample and hold control signal. A capacitor 230 isconnected between ground and the input terminal of the buffer amplifier216.

The buffer amplifier 222, the switch 223, and the capacitor 232 of thesecond sample and hold circuit 218 are correspondingly connected betweenthe delay circuit 219, the other terminal of the switch 225, the outputterminal 188 of the second receiver winding 179 and ground.

As shown in FIG. 8, the microprocessor 226 is connected to the gate of atransistor 210 of the signal generator 200. A supply voltage V+ isconnected through a bias resistor 212 to the drain of the transistor210. The source of the transistor 210 is connected to ground.

An LC series circuit is formed by a capacitor 214 and the transmitterwinding 180. The LC series circuit is connected between the drain of thetransistor 210 and ground. When the transistor 210 is off, the capacitor214 is connected through the resistor 212 to the supply voltage V+ andis charged to the supply voltage V+. The supply voltage is preferablysupplied by an appropriate power source (not shown), such as a battery.Together, the power supply V+, the transistor 210, the resistor 212 andthe capacitor 214 form the first preferred embodiment of the signalgenerator circuit 200 of the signal processing and display electroniccircuit 166.

To turn on the transistor 210, the microprocessor 226 supplies a shortpulse to the gate of transistor 210. When the transistor 210 is on, thecapacitor 214 is connected to ground through the transistor 210. Becausethe capacitor voltage cannot change instantly, the voltage at the node Abetween the capacitor 214 and the transmitter winding 180 is driven to anegative value.

Then, the capacitor 214 and the transmitter winding 180 resonate witheach other at a frequency determined by the capacitance of the capacitor214 and the inductance of the transmitter winding 180.

During each sampling period of the receiver output signal, the capacitor214 is discharged and then recharged. For the low power induced currenttransducer used in the first and second preferred embodiments of thelinear scale 100, a sampling frequency of about 1 kHz is preferred. Thecapacitor 214 preferably has a value of 1 nF and the power supplyvoltage V+ is preferably 3 V.

The charge (in coulombs) provided by the power supply voltage V+ isequal to the capacitance of the capacitor times the change in voltageacross the capacitor (coulomb=farad*volt). Accordingly, the chargestored by the capacitor 214 is equal to the capacitance of the capacitor214, 1 nF, times the voltage across the capacitor, 3 V, or 3 nC.

The capacitor will discharge and recharge every sampling period, whichis lms for a sampling rate of 1 kHz.

Additionally, current is charge divided by time (amp=charge/second).Accordingly, the average current drawn from the power supply during onesampling interval is 3 nC/1 ms=3 μA. Three microamps is a very smallcurrent, even for a battery-powered transducer.

In practice, the sampling frequency is preferably sufficiently high sothat the transducer does not move more than 1/4 of the wavelengthbetween measurements. For a wavelength of 0.2 inches (5.08 mm), asampling rate of 1 kHz and maximum allowable error during motion of 1/4wavelength (1.27 mm) the maximum allowable speed is 1.27 m/s. To obtainhigher speeds, or to reduce the maximum error at this speed, thesampling rate can be increased. Alternatively, the analyzing circuit caninclude a conventional second-order position follower, rather thanincreasing the sampling rate.

Because of the chosen circuit configuration, the signal generator 200enables the transducer to intermittently provide a strong output signal(about 60 mV maximum) across the terminals 185-188 of the receiverwindings 178 and 179. However, because of the inventive circuit andtransducer configuration, and because the driving signal is a shortpulse and operates with a low duty cycle, this results in a very smallaverage current, even at the desired rapid sampling rate of 1 kHz. Asmall average current is required for a battery or solar cell-poweredelectronic linear scale to be commercially practical.

To minimize power consumption, the input pulse applied to the transistor210 should be as short as possible, so that the charge lost through thebias resistor 212 is minimized. In the example outlined above, if thepulse length is 1 μs and resistor 212 has a value of 10 kΩ, the averagecurrent through the resistor 212 is only 0.3 μA. In general in thisinvention, the average current used to charge the capacitor 214 ispreferably less than 75 μA, and more preferably less than 10 μA. Asshown in FIG. 8, the capacitors 230 and 232 are electrically connectedin parallel to the receiver windings 178 and 179, respectively. Thecapacitances of the capacitors 230 and 232 form resonant circuits withthe inductances of the receiver windings 178 and 179. If the resonantfrequency of these resonant circuits is the same as the resonantfrequency of the transmitter resonant circuit, the strength of thesignal output from the receiver windings is increased and unwanted noiseis filtered from the signals.

Since the transmitter winding 180, as an inductor, and the capacitor 214form a resonant LC circuit, a voltage transient measured at node A willhave a decaying resonant behavior, as shown in FIG. 9. The transientvoltage signal causes a corresponding current flow in the transmitterwinding 180. This, in turn, produces a changing magnetic flux normal tothe loops 191 and 192 of the receiver windings 178 and 179,respectively.

The receiver windings 178 and 179 each have two conductor portions. Asshown in FIG. 8, these portions are placed at spatial phase positionscorresponding to 0° and 180° for the first receiver winding 178, and at90° and 270° for the second winding 179. As previously described, withthe scale 104 and the disrupters 170 in place, a current is induced inthe disrupters 170. The field from this induced current results in a netEMF in the receiver windings 178 and 179.

The degree notation in FIG. 8 (0°, 90°, 180°, 270°) corresponds to thelocation of the different parts of the windings 178 and 179 relative toa nominal position. The serial connection of the two halves of the firstreceiver winding 178, for example, causes the voltage at the output ofthe first receiver winding 178 to have one polarity when the disrupters170 are in the 0° position. The voltage at the output of the firstreceiver winding 178 has the opposite polarity when the disrupters 170are in the 180° position.

The signals from the receiver windings 178 and 179 are transmitted tothe signal processing and display electronic circuit 166. The signalprocessing and display electronic circuit 166 analyzes the signals todetermine the distance being measured by the linear scale 100. Thesignal processing and display electronic circuit 166 then is connectedto, and provides a drive signal to, the display 138 through aconventional connection to provide a digital readout of the measureddistance.

FIGS. 10A-10C show the voltage induced across the output of the firstreceiver winding 178 in response to the transmitter winding's transientvoltage excitation, as shown in FIG. 9. In particular, FIGS. 10A-10Cshow the induced voltage for three different positions of the disrupters170 relative to the loops 191 of the first receiver winding 178. Theamplitude and phase of the receiver signal depend on the position of thescale 104 relative to the receiver winding 178 or 179.

The receiver signal shown in FIG. 10A has a peak amplitude at point B.The peak amplitude indicates that the relative position between thescale and the receiver winding 178 or 179 is such that a maximumamplitude signal results. The inventors have experimentally determinedthat if the inductance of the transmitter winding 180 is 0.5 μH, thecapacitance of the capacitor 214 is 1 nF, the gap 174 is approximately0.5 mm, and the power supply voltage V+ is 3 V, then the maximumreceiver output signal at point B will be approximately 60 mV. The LCseries circuit formed by the capacitor 214 and the transmitter windinghas a resonant frequency of approximately 7 MHz.

FIG. 10B shows the receiver signal when the scale 104 is moved 1/4 ofthe wavelength 193 from the relative position generating the receiversignal shown in FIG. 10A. As shown in FIG. 10B, this relative positionbetween the first receiver winding 178 and the disrupters 170 generatesa receiver output signal having an amplitude of zero at point B. Thissignal corresponds to a position where each disrupter 170 overlaps equalareas of adjacent "+" loops 191a and "-" loops 191b for the receiverwinding 178.

In FIG. 10C, the scale has been moved another 1/4 of the wavelength 193in the same direction, for a total displacement of 1/2 of a wavelength193 from the relative position generating the receiver signal shown inFIG. 10A. In this relative position, the disrupter 170 overlaps a loop191 of opposite polarity from the loop 191 corresponding to FIG. 10A.Accordingly the first receiver winding 178 produces a maximum negativeamplitude receiver signal at point B.

FIGS. 11A-G show the signals present at various points of the signalprocessing and display electronic circuit 166. As shown in FIGS. 11D and11E, the disrupters 170 are positioned relative to the loops 191 and 192such that the receiver signals output by the receiver windings 178 and179 are equal and opposite. The equal and opposite receiver signalsoutput by the receiver windings 178 and 179 are input to the sample andhold circuits 217 and 218, respectively, of the signal processing anddisplay electronic circuit 166.

FIG. 11A shows the transmission control signal output by themicroprocessor 226 to the gate of the transistor 210. As shown in FIG.11A, the transmission control signal has a duration of t. FIG. 11B showsthe resulting oscillating transmitter signal applied to the transmitterwinding 180.

FIG. 11C shows the sample and hold control signal output from the delaycircuit 219. The delay circuit 219 inputs the transmission controlsignal shown in FIG. 11A from the microprocessor 226 to simultaneouslyinitiate the sample and hold control signal. The duration of the sampleand hold signal is chosen according to circuit and transducer designparameters, either by analysis or experiment. In particular, theduration is chosen such that the trailing edge of the sample and holdcontrol signal coincides as closely as possible in time with theamplitudes of the receiver signals reaching point B, as shown in FIGS.10A and 10C. In response to the sample and hold control signal outputfrom the delay circuit 219, the first and second sample and holdcircuits 217 and 218 sample the signals from the first and secondreceiver windings 178 and 179, respectively.

The sample and hold control signal closes the switches 221 and 223generally simultaneously with the start of the transmission signal beingapplied to the transmission winding 180. The signals output by the firstand second receiver windings 178 and 179 appear on the capacitors 230and 232, respectively, as shown in FIGS. 11F and 11G.

At a time nominally chosen to coincide with time B, the sample and holdcontrol signal returns to zero and the switches 221 and 223 open. Thevoltages across the capacitors 230 and 232 at that instant are thenheld. In general, the sampled voltage may be held at any time during thereceiver signals shown in FIGS. 11D and 11E, except at the zerocrossings. The time B is the preferred hold time and occurs when themaximum receiver signal strength is obtained.

As illustrated in FIGS. 11B and 11C, this corresponds to a peak of theresonant response. The time B is established by the delay circuit 219,which is, for example, a monostable flip-flop triggered by thetransmission control signal.

The sampled voltages are input to the high input impedance bufferamplifiers 216 and 222. The buffer amplifiers 216 and 222 provide gainand isolate the capacitors 221 and 223 to prevent the capacitors 221 and223 from losing their charge. The buffer amplifiers 216 and 222,respectively, output a signal S₁, corresponding to the receiver signaloutput by the receiver winding 178, and a signal S₂, corresponding tothe receiver signal output by the receiver winding 179. The selectorswitch 225 alternately couples the outputs of the buffer amplifier 216or 222 to the A/D converter 224. The A/D converter 224 converts theanalog signals S₁ and S₂ to digital signals.

The microprocessor 226 inputs the digital signal from the A/D converter224, computes a measurement position, and outputs appropriate signals tothe display 138. The microprocessor 226 can evaluate the position of thescale 104 in any one of a variety of methods, including the methoddefined by Equation (1).

The display button logic, the system control logic, analysis ofdisplacements exceeding one wavelength, and other typical electronicscale functions are preferably provided as in the prior art capacitiveelectronic linear scales, such as those produced by Mitutoyo, Brown &Sharp, Sylvac, Starret, etc. The preferred embodiment displays themeasured distance on the display 138. The computed measurement positioncan also be output to other systems through suitable connections (notshown) similar to those in commercially available capacitive linearscales. For example, the computed measurement data may be output to astatistical process control system or to a remote measurement display.

The signal processing and display electronic circuit 166 can be readilyincorporated into the linear scale 100 by mounting the elements of thesignal processing and display electronic circuit 166 on the substrate162. In some cases, a conventional multi-layer printed circuit substratecan be used, so that the internal layers of the substrate can be used toprovide conventional ground-plane shielding (not shown) betweennoise-generating and sensitive parts of the signal processing anddisplay electronic circuit 166. This eliminates unwanted interactionbetween the electronic signals in these elements.

As shown in FIGS. 11A-G, the transmission control signal remains highfor several peaks of the resonant response. However, as shown in FIGS.12A-G, the transistor 210 may be turned off after a sufficient period toallow sampling of the capacitor voltages. The transistor 210 does notneed to remain on beyond the sampling time.

Thus, to conserve power, the transistor 210 can be turned off before theresonant circuit has dissipated its stored energy. Preferably, as shownin FIG. 12A, the transistor 210 is turned off at time C when the voltageacross the capacitor 214 has returned as close as possible to itsoriginal value. In the preferred embodiment described above, theoriginal value is the battery voltage V+.

It should also be appreciated that sufficient time must be providedbetween successive transmission control pulses, in order to allow thecapacitor 214 to fully recharge. Generally, if the circuit comprisingthe capacitor 214 and the resistor 212 has a time constant T_(c), thetime allowed between successive transmission control pulses, i.e., thepulse interval of the drive signal, should be at least 4 times the timeconstant T_(c).

A second preferred, and low-power, embodiment of the signal generator200 is shown in FIG. 13. In the first preferred embodiment of the signalgenerator 200, energy is lost through the bias resistor 212 when thetransistor 210 is turned on. The second preferred embodiment of thesignal generator 200 eliminates a substantial portion of this energyloss by using an active pull up switch 240 in place of the bias resistor212 to bias the transistor 210. This energy loss is minimized becausethe resistance of the pull up switch 240 when it is open, is muchgreater than the resistance of the bias resistor 212.

The active pull-up switch 240 also allows the capacitor 214 to becharged much faster. Thus, the low on-resistance of the switch 240allows a much higher sampling rate compared to the first preferredembodiment of the signal generator shown in FIG. 8. In the secondpreferred embodiment of the signal generator 200, the switch 240 and thetransistor 210 are controlled by a pair of synchronous control signals.

The microprocessor 226 generates the switch control signal input to thecontrol switch 240 and the transistor 210. The switch control signal tothe switch 240 is high when the transmission control signal to thetransistor 210 is low. In this condition, the switch 240 is closed andthe capacitor 214 is charged to V+ through the transmission winding 180.The transmission control signal to the transistor 210 is low, turningoff the transistor 210.

This arrangement prevents the switch 240 and the transistor 210 fromconducting at the same time. This prevents a large current draw from thebattery and preserves battery life.

When the capacitor 214 is charged, the control switch signal opens theswitch 240. Thereafter, the transmission control signal to thetransistor 210 turns on the transistor 210. It should be appreciatedthat the transistor 210 remains off until after the switch 240 is off sothe switch 240 and the transistor 210 do not conduct at the same time.

When the transistor 210 is turned on, the is capacitor 214 is connectedto ground. The charged capacitor 214 and the transmitting winding 180form a resonant circuit. Because the capacitor 214 is charged, thevoltage across the transmitter winding 180 resonates as shown in FIG. 9.

The corresponding current through the transmitter winding also generatesthe changing magnetic field through the receiver windings 178 and 179.The disrupters 170 then induce a net signal in the receiver windings 178and 179. The delay circuit 219 controls the sample and hold circuit inrelation to the transmission signal to transistor 210, as previouslydescribed. The amplitude and sign of the net signal will depend upon theposition of the disrupters 170 relative to receiver windings 178 and179.

A high Q-value corresponds to low energy loss in the resonant circuit. Ahigh Q-value is desirable because the voltage across the capacitor 214swings back closer to the battery voltage V+. Thus, if the transmissioncontrol signal is turned off at time C, as shown in FIGS. 12A and 14,the voltage across the capacitor 214, V_(peak), will be only slightlybelow the battery voltage V+. Thus, the battery needs to provide only asmall amount of charge to replenish or recharge the capacitor 214 inpreparation for the next transmission/reception cycle.

As shown in FIGS. 12F and 12G, turning the transmission control signaloff at time C does not affect the sampled signals on the capacitors 230and 232 because the transistor 210 remains on until after the samplingtime B.

It is also possible to operate the linear scale transducer in reverse,i.e., to transmit through the windings 178 and 179 and receive, or sensemagnetic flux, through winding 180. The encoder electronics for thismode of operation are shown in FIG. 15. FIG. 16 is a signal timingdiagram showing the timing of the control signals.

As described below and shown in FIG. 16, it is possible to firsttransmit a signal from one of the windings 178 and 179, receive thetransmitted signal through winding 180, and then transmit a secondsignal from the other one of the windings 178 and 179 to the winding180, to obtain information necessary to determine the location of thetransducer on the linear scale.

Although it is possible to have two signal generators (not shown), inpractice it is difficult to manufacture generators that are sufficientlywell matched because of variations in the discrete electrical componentswithin the generators. Furthermore, although signals can besimultaneously broadcast from two transmitters (e.g. windings 178 and179) and received through the winding 180, using such information todetermine location of the of the transducer on the linear scale wouldrequire that the signal processing include phase analysis, which isbeyond the scope of this application. Accordingly, the process describedbelow refers to a signal generator 200 that causes a first signal to betransmitted from one of the windings 178 and 179, and then subsequentlya second signal from the other one of the windings 178 and 179.

The microprocessor 226 controls three selector switches 324, 326, and328 with the selector control signal. When the selector control signalgoes high, the switches 326 and 328 are moved to the positions shown inFIG. 15. In these positions, the winding 178 is connected to the signalgenerator and the winding 180 is connected to the sample-and-holdcircuit 217. The sample and hold control signal is also routed tosample-and-hold circuit 217. Next, the microprocessor 226 outputs atransmission control pulse to the signal generator 200 and to the delaycircuit 219. The signal S₁ is thus sampled and held by the switch 221and the capacitor 230.

The microprocessor 226 then changes the selector control signal to low.This moves switches 326 and 328 to their alternate positions. Thus, thewinding 179 is connected to the signal generator 200 and the winding 180is connected to the sample-and-hold circuit 218. The sample and holdcontrol signal is also routed to the sample-and-hold circuit 218. Themicroprocessor 226 outputs a new transmission control signal to thesignal generator 200 and to the delay circuit 219. The signal S₂ is thussampled and held by the switch 223 and the capacitor 232. The rest ofthe signal processing is the same as described in conjunction with FIG.8.

In the encoder electronics shown in FIG. 15, a single signal generatoris used, and is alternately connected to the transmitter windings 178and 179. It is also possible to use two signal generators, eachconnected to one of the transmitter windings 178 and 179.

The microprocessor 226 is able to determine an absolute positionmeasurement within one-half of the wavelength 193, using knowninterpolation routines and only one receiver winding. For example, asshown in FIG. 17, the microprocessor 226 can distinguish between a firstposition d1 and a second position d2 within one-half of the wavelength193 by comparing the amplitude and polarity of the receiver signal atpoints 387 and 388, respectively.

The point 387 has a voltage value of V1, while point 388 has a voltagevalue of V2. The position d3 corresponds to the point 389 in thereceiver signal shown in FIG. 17. The point 389 has the same voltagevalue V1 as the point 387. Therefore, the microprocessor 226 cannotdetermine the difference in relative position between the first positiond1 and the third position d3 using the interpolation approach.

In the preferred embodiments described above, the microprocessor 226resolves such ambiguities by using the signal from the second receiverwinding 179 according to well-known quadrature signal analysistechniques, as shown in Eq. 1. For motion beyond one wavelength, themicroprocessor 226 detects and accumulates the number of wavelengthstraversed from a known starting position, according to well-knowntechniques, to determine the relative positions of the read head 164 andthe scale 104.

The microprocessor 226 supplies pulses at a sampling frequency of about1 kHz to provide sufficient accuracy and motion tracking capability. Toreduce power consumption, the microprocessor 226 also keeps the dutycycle low by making the pulses relatively short. For example, for the 1kHz sampling frequency described above, typical pulse width is about0.1-1.0 μs. That is, the duty cycle of the pulses having sampling periodof 1 ms is 0.01%-0.1%.

The resonant frequency of the capacitor 214 and the winding 180 is thenpreferably selected such that the peak of the voltage across thecapacitor 214 occurs before the end of the at most 1.0 μs pulse. Thus,the resonant frequency is on the order of several megahertz. Thecorresponding magnetic flux will therefore be modulated at a frequencyabove 1 MHz, and typically at a frequency of several megahertz. This isconsiderably higher than the frequencies of conventional inductivetransducers.

The inventors have determined that, at these frequencies, the eddycurrents generated in the disrupters 170 produce a strong disruptiveeffect on the magnetic flux. The output EMFs of the receiver windings178 and 179 therefore respond strongly to variations in disrupterposition. This occurs despite the low duty cycle and low power used bythe pulse signal. The strength of the response, combined with the lowduty cycle and low power consumption, allows the linear scale 100 tomake measurements while the signal generator 200 and the remainder ofthe signal processing and display electronic circuit 166 draw an averagecurrent preferably below 200 μA, and more preferably below 75 μA. Itshould be understood that "average current" as used herein means thetotal charge consumed over one or more measurement cycles, divided bythe duration of the one or more measurement cycles, while the caliper isin normal use.

The linear scale 100 can therefore be operated with an adequate batterylifetime using three or fewer commercially available miniature batteriesor with a solar cell.

For an incremental-type transducer, the rate at which the transducersignals are sampled is about 1000 samples per second. The high samplingrate is required to keep track of the number of wavelengths traveled,when the slider assembly 120 is moving quickly. However, themicroprocessor 226 only needs to update the display 138 with a newmeasurement value about 10 times per second. Therefore, the powerconsumption of the linear scale can be reduced further if themicroprocessor 226 and A/D converter 224 can be relieved of the task ofperforming high resolution position measurements for a majority of the1000 samples per second. This can be accomplished by keeping track ofthe number of scale wavelengths traveled without performing highresolution position measurements.

FIG. 18 shows an example of encoder electronics where a wavelengthtracker 320 keeps track of the wavelength number. The wavelength trackerconsumes very little power. With a wavelength tracker, themicroprocessor 226 can go into a sleep mode when it is not performing ahigh resolution position measurement and thus not updating the display138, thereby saving power. A transmit/receive sequencer 322 generatesthe control signals for the signal generator 200 and the sample-and-holdcircuits 217 and 218 by conventional circuit means. The transmit/receivesequencer 322 also generates a strobe pulse for the wavelength tracker320 and outputs a display update control signal to the microprocessor226.

The wavelength tracker 320 includes two analog comparators 310 and 312,a quadrature counter 318, and a control logic unit 314. The comparators310 and 312 detect the zero crossings of the signals S₁ and S₂. Areference voltage V_(ref) is input to each of the comparators 310 and312. The output states of the comparators are read into a quadraturecounter 318 on command from the wavelength tracker strobe when theoutputs of the comparators have stabilized. The quadrature counter 318counts the number of full wavelengths traveled. The quadrature counter318 is an up/down counter. The quadrature counter 318 detects thedirection of movement of the slider assembly 120 because the signals S₁and S₂ are in quadrature. That is, because these signals are phaseshifted 90° relative to each other. The quadrature counter 318 is acircuit well known in the art. Quadrature counters are, for instance,commonly used to detect the position of optical rotary encoders andlinear scales.

Where the integer wavelength count of the wavelength tracker 320 differsfrom that of the microprocessor 226, the electronic linear scale usesthe count of the wavelength tracker 320 rather than of themicroprocessor 226. For measurements or increments less than awavelength however, the microprocessor has priority.

The control logic unit 314 shifts the microprocessor 226 between sleepand active modes at appropriate times, for example when the sliderassembly 120 is moving quickly as described above. The control logicunit 314 also can be used to shift the microprocessor 226 from theactive mode to the sleep mode when the slider 120 has not moved for aspecified time period. Subsequently, the control logic unit 314 can alsobe used to shift the microprocessor 226 from the sleep mode to theactive mode when the slider 120 begins to move. Those skilled in the artwill recognize other ways in which the control logic unit 314 can beused to appropriately shift the microprocessor 226 between active andsleep modes in order to reduce power consumption.

Debounce logic (not shown) is also included within the wavelengthtracker to prevent erroneous measurements due to edge jitter when theslider 120 is near a wavelength or quadrature transition. Such debouncelogic is well known, and apparent to those of ordinary skill in the art.

FIG. 19 shows a signal timing diagram of the signals output from thetransmit/receive sequencer 322, and associated signals. FIG. 20 showsthe transmission control signal and the display update control signal.

When the microprocessor 226 updates the display 138 (for example, 10times per second) it calculates the number of wavelengths traveled fromthe "zero" position. It reads the number of full wavelengths traveledfrom the quadrature counter 318. The microprocessor 226 then calculatesthe fractional wavelength traveled based on the signals S₁ and S₂. (Thesignals S₁ and S₂ have earlier been converted to digital signals by theA/D converter 224.) The fractional wavelength is added to the number offull wavelengths and the result is multiplied by the wavelength toobtain the position value fed to the display.

The transmit/receive sequencer 322 controls sampling of the transducersignals and the wavelength tracker counts the number of wavelengthstraveled, without any help from the A/D converter 224 or microprocessor226. In this embodiment, the linear scale is made "quasi-absolute" byturning off only the microprocessor 226, the A/D converter 224, and thedisplay 138 when the linear scale is turned off, so that the samplingcircuits and the wavelength tracker 318 are still active. While thelinear scale was turned off, the wavelength tracker remained active.Thus, the wavelength tracker provides information regarding thewavelength at which the linear scale is positioned, even if the positionwas moved during the time the linear scale was turned off. Thus, whenthe linear scale is turned back on, the position of the linear scale canbe computed and displayed with reference to the original "zero"position, despite the fact that the high resolution measurement anddisplay functions of the linear scale were turned off.

FIGS. 21 and 22 show a third preferred embodiment of the electroniclinear scale 100 of this invention.

In this third preferred embodiment, the disrupters 170 are integrallyformed in a conductive beam 102', as shown in FIG. 22. The upper surfaceof the conductive beam 102' is etched or machined to form evenly spacedgrooves 220, leaving raised portions of the conductive beam 102'. Theupwardly projecting raised portions of the conductive beam 102' thusform the disrupters 170. An insulative layer 172 is formed over theupper surface of the conductive beam 102' and covers the disrupters 170and the grooves 220. An air gap 174 is provided between the insulativecoating 167 of the read head 164 and the insulative layer 172. The sizeof the air gap 174 is preferably the same as in the first preferredembodiment, i.e., on the order of 0.5 mm.

As shown in FIG. 21, power for operating the linear scale 100 can bedrawn from a conventional solar cell 227 mounted in the cover 114, withpower routed to the signal processing electronics 166 by conventionalmeans (not shown). A transparent protective cover for the solar cell(not shown) is preferred. The conventional solar cell 227 is acommercially available component and supplies adequate power to operatethe low power inductive transducer linear scale 100. Alternately, thesolar cells can be mounted on digital display unit 139. This removesthem from the higher contaminant concentration around the sliderassembly 120.

The first, second and third preferred embodiments of the linear scale100, as described above, include flux disrupters 170 to interact withthe read head 164 to provide the measurement signal. In a fourthpreferred embodiment, as shown in FIGS. 23 and 24, flux enhancers 170'are used in place of the flux disrupters 170. A flux enhancer 170'"enhances" or increases the magnetic flux through the adjacent portionsof the receiver windings 178 and 179.

Except as noted below, the fourth preferred embodiment of the linearscale 100 shown in FIGS. 23 and 24 can use any of the read headgeometries, circuits and mechanical configurations disclosed in thefirst, second or third preferred embodiments. In all cases, it should beunderstood that when a flux enhancer 170' is used in place of a fluxdisrupter 170, the magnetic field will be enhanced rather thandisrupted.

That is, the flux density is increased and the polarity of the resultingsignals will be inverted when the flux enhancers 170' are used, relativeto the effects generated when the flux disrupters 170 are used. Ineither case, the flux enhancer 170' or flux disrupter 170 spatiallymodulates the magnetic flux.

The enhancement-type linear scale 100 shown in FIGS. 23 and 24 enhancesthe magnetic flux by moving an object of high magnetic permeability,such as ferrite, close to the read head 164. The flux enhancers 170'provide a lower reluctance path for the varying magnetic field generatedby the transmitter winding 180. As a result, the magnetic flux that thereceiver windings 178 and 179 receive is altered or enhanced in thevicinity of the flux enhancers 170'. This causes the receiver windings178 and 179 to output non-zero EMF signals.

Consequently, if the flux enhancers 170' each have a length equal to 1/2of the wavelength 193, the signal measured at the output terminals185-188 of the receiver windings 178 and 179 will change polarity andamplitude as the flux enhancers 170' move between the "+" and "-" loops191a and 191b of receiver winding 178 and 192a and 192b of receiverwinding 179. Thus, the enhancement-type linear scale 100 of the fourthpreferred embodiment operates in a manner completely analogous to theabove described signal behavior resulting from the disrupters 170 usedin the first, second and third preferred embodiments of the linear scale100.

If high permeability objects such as the enhancers 170' are movedrelative to the read head 164, the regions of higher flux densitythrough the receiver windings 178 and 179 interact with successive onesof the loops 191 and 192. The AC amplitude of the signal output from thereceiver windings 178 and 179 will depend upon the difference betweenthe area of the "+" loops 191a and 192a overlapped by the flux enhancers170' and the area of the "-" loops 191b and 192b overlapped by the fluxenhancers 170'.

As the flux enhancers 170' travel along the measuring axis 300, the ACamplitudes of the signals output from the receiver windings 178 and 179vary continuously due to the continuous relationship between theoverlapped areas of the "+" loops 191a and 192a and the overlapped areasof the "-" loops 191b and 192b. The signals will also vary periodicallywith the wavelength 193 due to the periodically alternating "+" and "-"loops 191a and 191b of the receiver winding 178 and 192a and 192b ofreceiver winding 179, as shown in FIG. 4, and the dimension andplacement of the flux enhancers 170'.

The signals output from the receiver windings 178 and 179 have smooth,continuous, sinusoidal shapes based on the movement of the fluxenhancers 170' relative to the receiver windings 178 and 179. Continuoussignals enable the linear scale 100 to make accurate position readingsover extended distances.

In the fourth preferred embodiment of the linear scale 100 shown inFIGS. 23 and 24, the scale 104 includes multiple flux enhancers 170'positioned on and spaced apart along the substrate 168'. The fluxenhancers 170' are rectangular members having high magneticpermeability. The flux enhancers 170' are preferably formed from anon-conductive, e.g., highly resistive material, such as ferrite. Theflux enhancers 170' are also non-magnetized, so that they do not attractferromagnetic particles.

The substrate 168' is preferably formed from a material having asubstantially lower magnetic permeability than the material of the fluxenhancers 170'. Similarly to the flux disrupters 170, the flux enhancers170' preferably have a length equal to one-half of the wavelength 193and are arranged at a pitch equal to one wavelength 193. The thicknessof the flux enhancers 270' is preferably on the order of 1.5 mm. Theresulting signal strength is comparable to that of a disrupter-typelinear scale 100.

Although the flux enhancers 170' can be thicker or thinner than 1.5 mm,thicker flux enhancers 170' will provide a greater signal strength. Theactual thickness of the flux enhancers 170' will be determined by thetradeoff between the desired signal strength and the material andmanufacturing costs.

The substrate 168' is preferably non-conductive, as in the first andsecond preferred embodiments. However, the substrate 168' may be more orless conductive depending on manufacturing considerations. The fluxenhancers 170' are shown in FIGS. 23 and 24 as formed from a materialseparate from the beam 102 and the substrate 168'. However, as in thethird preferred embodiment of the linear scale 100, the flux enhancers170' can be integrally formed with the beam 102'. In this case, the fluxenhancers 170' will be formed by processes that alter the permeabilityof the portions of the material of the beam 102' that form the fluxenhancers 170'.

As shown in FIG. 25, in a fifth preferred embodiment of the linear scale100, the flux enhancers 170' are formed as raised or protruding portionsof the beam 102'. Preferably a surface contouring process is used toform the protruding portions. Thus, as in the third preferred embodimentof the linear scale 100 shown in FIG. 22, flux modulators are integrallyformed from the same material as the beam 102'. However, in this case,they are flux enhancers 170'.

The close proximity of the flux enhancers 170' to the read head 164decreases the magnetic path reluctance for the magnetic flux in thevicinity of the flux enhancers 170'. This effect is comparable to thepermeability variation between the substrate 168', the flux enhancers170' and the empty spaces in the fourth preferred embodiment shown inFIGS. 23 and 24. This allows the fifth preferred embodiment of thelinear scale 100 to operate substantially similarly to the fourthpreferred embodiment of the linear scale 100 of FIGS. 23 and 24.

In a sixth preferred embodiment of the linear scale 100, as shown inFIG. 26, the beam 102' (or the substrate 168') includes a plurality ofless magnetically permeable segments 233, such as alumina, alternatingwith a plurality of high-permeability, highly resistive segments 234,such as ferrite. The beam 102' or the substrate 168' is thus formed bythe series of alternating segments 233 and 234 bonded to form analternating stack of material. The relatively more magneticallypermeable, non-conductive segments 234 define the flux enhancers 170'and provide a lower reluctance path than the less magnetically permeablesegments 233.

It should be appreciated that the less magnetically permeable segments233 can be formed from a conductive material, for example, copper orbrass. In this case, the less magnetically permeable segments are alsodisrupters 170. Therefore, the beam 102' or the substrate 168' includestwo types of flux modulators: the flux disrupters 233' and the fluxenhancers 234.

Thus, the flux enhancers 170' (234) and the flux disrupters 170 (233')can be alternately placed along the surface of the beam 102' or thesubstrate 168', as shown in FIG. 27. In this seventh preferredembodiment of the linear scale 100, the effects on the receiver signalscaused by the disrupters 170 (233') and the enhancers 170' (234) will beroughly additive, producing a stronger signal than when either type offlux modulator is used alone.

FIG. 28 shows the flux disrupters 170 (233') and the flux enhancers 170'(234) can be provided on a base 102' or on the substrate 168'.Additionally, as in the third or fifth embodiments, the flux disrupters170 (233') or the flux enhancers 170' (234) can be provided integrallywith the base 102' or the substrate 168', as shown in FIGS. 29 and 30.In this case, the other of the flux disrupters 170 (233') and the fluxenhancers 170' (234) are inserted into the grooves 220.

All of the geometric design principles and circuits disclosed above forthe flux-disrupter type linear scales of the first, second and thirdpreferred embodiments can of course be used with the flux-enhancementtype linear scales 100 of the fourth-seventh preferred embodiments toproduce the high accuracy and other benefits the first-third preferredembodiments of the linear scale 100 provide. The various read headgeometries described above, as well as the circuits and mechanicalconfigurations disclosed above, can all be employed to providesubstantial accuracy improvements to prior art "enhancer-type encoders"when the flux enhancers 170' are substituted for the flux disrupters170. The low-power circuit techniques described above can also be usedwith the flux enhancers 170' while retaining their low-power benefits.

Although specific embodiments of, and examples for, this invention havebeen illustratively described, various equivalent modifications can bemade without departing from the spirit and scope of this invention. Forexample, while sinusoidal loops 191 and 192 are shown and described withrespect to the receiver windings 178 and 179, various other geometriescan be effectively used, including different geometries for differentphases in a given read head.

Similarly, while generally rectangular conductive bars and rectangularhigh permeability bars are shown and described herein as two types ofspatial flux modulators, other geometries can be used. When thesegeometries lead to non-sinusoidal output signals as a function ofdisplacement, then the actual function can be modeled in a look-uptable, or by other means known to those skilled in the art. The positioncalculating equations described herein can be similarly modified orreplaced according to well-known signal processing techniques.

Those skilled in the art will also recognize that the sampling frequencycan be chosen to be higher or lower than that described above, dependingupon the desired accuracy and a maximum expected rate of change in thedistance being measured.

Also, the signal processing and display electronic circuit 166 describedherein includes only exemplary analyzing circuits. One skilled in theart will recognize that other circuits may be used to drive thetransmitter winding 180 and to detect signals from the receiver windings178 and 179. Also, one skilled in the art will recognize that, due tothe symmetry of the electromagnetic principles disclosed above, theoperating role of transmitter winding 180 and receiver windings 178 and179 can be reversed, as outlined above.

Those skilled in the art will further recognize that the electroniccomponents handling the high frequency signals are preferably located asclose as possible to the transducer, while the electronics handling thelow frequency signals can be more remotely located from the transducer.The high frequency electronics include, for example, the circuits usedto drive the transmitter windings and to detect the signals from thereceiver winding. The low frequency electronics include, for example,those circuits downstream of the sample and hold circuit. In particular,when the transducer excitation frequency is lMHz or greater, at leastthe signal generating circuits and the demodulating circuits should bepositioned very close to (or optionally, on) the read head 164.Likewise, locating the sample and hold circuitry on the read head 164may be desirable.

Accordingly, the present invention is not limited by the disclosure, butinstead, its scope is to be determined entirely by reference to thefollowing claims.

What is claimed is:
 1. An electronic inductive linear scale comprising:afirst member; an elongated beam having a measuring axis, the firstmember movable relative to the elongated beam along the measuring axis;a low power energy supply source providing a power supply to a drivecircuit; the drive circuit inputting the power supply and outputting anintermittent drive signal; an inductive transducer comprising a firstportion mounted on one of the elongated beam and the first member and asecond portion mounted on the other of the elongated beam and the firstmember, the inductive transducer inputting the intermittent drive signaland outputting at least one sensed signal responsive to a relativeposition of the first member on the elongated beam; and an analyzingcircuit inputting the at least one sensed signal and outputting anoutput signal responsive to the position of the first member on theelongated beam at a first level of resolution.
 2. The electronic linearscale of claim 1, wherein the drive circuit comprises a capacitordischarged through the inductive transducer.
 3. The electronic linearscale of claim 2, wherein the capacitor and the inductive transducerform a resonant circuit.
 4. The electronic linear scale of claim 3,wherein the capacitor is disconnected from the inductive transducer inclose proximity to a resonant peak of the resonant circuit.
 5. Theelectronic linear scale of claim 1, wherein the analyzing circuitcomprises a counter for counting fractions of cycles of the at least onesensed signal outputted from the inductive transducer in response tomotion of the first member along the elongated beam at a second level ofresolution coarser than the first level of resolution, the counterproviding an approximate relative position of the first member on theelongated beam.
 6. The electronic linear scale of claim 5, wherein thecounter is responsive at spatial intervals of at most 1/4 cycle.
 7. Theelectronic linear scale of claim 1, wherein:the first portion of theinductive transducer comprises:at least one magnetic field generator,each magnetic field generator generating a changing magnetic flux in aflux region in response to the drive signal, and at least one magneticflux sensor, each magnetic flux sensor positioned within the flux regionand sensing the magnetic flux in the flux region, each magnetic fluxsensor generating one of the at least one sensed signals; and the secondportion of the inductive transducer comprises at least one fluxmodulator, each flux modulator positionable within the flux region andcapable of altering a magnetic flux within a modulation region proximateto the flux modulator; wherein each sensed signal is indicative of arelative position between the magnetic flux sensor and the at least oneflux modulator based on the sensed magnetic flux.
 8. The electroniclinear scale of claim 7, wherein at least one of a) each magnetic fluxsensor, and b) each magnetic field generator, is formed in analternating pattern of polarity regions.
 9. The electronic linear scaleof claim 8, wherein the alternating pattern of polarity regionscomprises sinusoidally shaped areas bounded by conductive elements. 10.The electronic linear scale of claim 7, wherein in the absence of the atleast one flux modulator, the output signal generated by each magneticflux sensor is insensitive to the changing magnetic flux generated byeach magnetic field generator.
 11. The electronic linear scale of claim7, wherein each of the at least one flux modulator comprises one of a) aflux disrupter and b) a flux enhancer.
 12. The electronic linear scaleof claim 7, wherein each magnetic field generator includes a fieldgenerating conductor and each magnetic flux sensor includes a sensingconductor, the field generating conductor of each magnetic fieldgenerator and the sensing conductor of each magnetic flux sensorpositioned within a thin zone.
 13. The electronic linear scale of claim7, wherein each magnetic field generator and each magnetic flux sensorform a continuous spatially modulated inductive coupling.
 14. Theelectronic linear scale of claim 1, wherein a magnetic field of theinductive transducer changes at a rate equivalent to an oscillationfrequency of at least 1 MHz in response to the intermittent drivesignal.
 15. The electronic linear scale of claim 1, wherein theintermittent drive signal comprises at least one pulse signal.
 16. Theelectronic linear scale of claim 1, wherein:the at least one pulsesignal comprises a train of pulse cycles; and the analyzing circuitdetermines changes in the relative position at a coarse level ofresolution during each pulse cycle, and determines the relative positionat a finer level of resolution once during a plurality of the pulsecycles.
 17. The electronic linear scale of claim 1, wherein theelectronic linear scale draws at most an average current of 200 μA. 18.The electronic caliper of claim 1, wherein the energy supply source is aself-contained, low-power energy supply source.
 19. The electroniclinear scale of claim 18, wherein the low-power energy supply source isat least one of a battery and a solar cell.
 20. The electronic inductivelinear scale of claim 1, wherein energy is conserved by creating aresonant circuit between a capacitor and a magnetic field generator ofthe inductive transducer and disconnecting the capacitor in closeproximity to a resonant peak of the resonant circuit.
 21. The electronicinductive linear scale of claim 1, wherein an average current of at most75 μA is supplied from the power supply to the drive circuit.
 22. Amethod for operating a linear scale, the linear scale comprising:a scalemember having a measurement axis, the scale member mountable on a firstmounting member; a first member mountable on a second mounting member sothat the first member is adjacent to the scale member, the firstmounting member and the second mounting member movable relative to eachother along the measurement axis; a magnetic flux sensor provided on oneof the first member and the scale member; a flux modulator provided onthe other of the scale member and the first member; a magnetic fluxgenerator provided on one of the first member and the scale member,wherein the magnetic field generator and the magnetic flux sensor forman inductive coupling; a low-power energy supply source; and a drivecircuit, the method comprising:providing a low-power supply signal fromthe low-power energy supply source to the drive circuit; intermittentlyoutputting a drive signal to the magnetic flux generator; producing achanging magnetic flux with the magnetic flux generator in a flux regionin response to the intermittent drive signal; moving the first mountingmember relative to the second mounting member to a measurement positionwhich is relative to an established reference position, wherein the fluxmodulator modulates the inductive coupling between the first member andthe scale member corresponding to a relative position between the firstmounting member and the second mounting member; sensing the modulatedmagnetic flux with the magnetic flux sensor to produce a sensed signalcorresponding to the relative position; monitoring the sensed signalproduced by the magnetic flux sensor; and determining, in response tothe monitored sensed signal, a distance between the establishedreference position and the measurement position.
 23. The method of claim22, wherein the magnetic flux generator is a transmitter winding and thestep of producing a changing magnetic flux comprises providing a drivingsignal to the transmitter winding, the driving signal causing a changingcurrent to flow through the transmitter winding to induce the changingmagnetic flux.
 24. The method of claim 23, wherein the step of providinga driving signal to the transmitter winding comprises:producing a seriesof pulses at a selected pulse interval with a pulse generator to producea pulsed signal; and supplying the pulsed signal to an input terminal ofthe transmitter winding.
 25. The method of claim 24, wherein the step ofmonitoring the sensed signal produced by the magnetic flux sensorincludes sampling the sensed signal synchronously with the pulsedsignal.
 26. The method of claim 25, wherein the step of sampling thesensed signal synchronously with the pulsed signal includes sampling thesensed signal based on an expected time delay between the pulsed signaland a peak in a response of a resonant circuit formed by the pulsegenerator and the transmitter winding.
 27. The method of claim 24,wherein the determining step comprises:determining changes in relativeposition at a coarse level of resolution during each pulse interval; anddetermining the distance between the established reference position andthe measurement position at a finer level of resolution once during aplurality of pulse intervals.
 28. The method of claim 22, furthercomprising the step of operating the electronic linear scale at at mostan average current draw of 200 μA.
 29. The method of claim 22, whereinthe step of producing a changing magnetic flux with the magnetic fluxgenerator includes supplying an average current at at most 75 μA to themagnetic flux generator.
 30. The method for operating an electroniclinear scale of claim 22, wherein the magnetic flux generator and themagnetic flux sensor form a continuous spatially modulated inductivecoupling, such that, when moving the first mounting member, the fluxmodulator further modulates the modulated inductive coupling between theslide member and the beam member to provide a continuously changingmagnetic flux.
 31. The method for operating the linear scale of claim22, wherein the energy supply source is a self-contained, low-powerenergy supply source.
 32. The method for operating the linear scale ofclaim 22, wherein the low-power energy supply source is at least one ofa battery and a solar cell.
 33. An electronic inductive linear scalecomprising:a first member; an scale member having a measuring axis, thefirst member movable relative to the scale member along the measuringaxis; at least one magnetic field generator, each magnetic fieldgenerator generating a changing magnetic flux in a flux region inresponse to a drive signal; a low-power energy supply source outputtinga power supply; a drive circuit inputting the power supply andoutputting an intermittent drive signal to at least one of the at leastone magnetic field generator; at least one flux modulator, each fluxmodulator positionable within the flux region and capable of alteringthe changing magnetic flux within a modulation region proximate to theflux modulator; and at least one magnetic flux sensor, each magneticflux sensor positioned within the flux region and sensing the changingmagnetic flux in the flux region, each magnetic flux generatorgenerating an output signal indicative of a relative position betweenthe magnetic flux sensor and the at least one flux modulator based onthe sensed magnetic flux; wherein each magnetic field generator and eachmagnetic flux sensor from a spatially modulated inductive couplingincluding an alternating pattern of polarity regions, and each outputsignal varies as a substantially linear function of a total overlappingarea, the overlapping area defined by a maximum cross-sectional area ofeach at least one flux modulator which overlaps both the flux region andan effective area of each at least one magnetic flux sensor whenprojected normal to an effective plane of the at least one flux sensor,where areas of opposite inductive coupling polarity are defined as areasof opposite sign.
 34. The electronic linear scale of claim 33, whereinthe area of at least one of a)each magnetic field generator and b)eachmagnetic flux sensor is sinusoidally modulated at a chosen spatialfrequency to define the continuously spatially modulated inductivecoupling.
 35. The electronic linear scale of claim 33, wherein the totaloverlapping area varies as a sinusoidal function at a chosen spatialfrequency as a function of the relative position.
 36. The electroniclinear scale of claim 33, wherein the spatially modulated inductivecoupling is a continuous spatially modulated inductive coupling.